AN1042 Datasheet PDF - ON Semiconductor
| Part Number | AN1042 | |
| Description | High Fidelity Switching Audio Amplifiers Using TMOS Power MOSFETs | |
| Manufacturers | ON Semiconductor | |
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AN1042/D
High Fidelity Switching
Audio Amplifiers Using
TMOS Power MOSFETs
Prepared by: Donald E. Pauly
ON Semiconductor
Special Consultant
http://onsemi.com
APPLICATION NOTE
Almost all switching amplifiers operate by generating a
high frequency square wave of variable duty cycle. This
square wave can be generated much more efficiently than
an analog waveform. By varying the duty cycle from 0 to
100%, a net dc component is created that ranges between
the negative and positive supply voltages. A low pass filter
delivers this dc component to the speaker. The square wave
must be generated at a frequency well above the range of
hearing in order to be able to cover the full audio spectrum
from dc to 20 kHz. Figure 1 shows a square wave
generating a sine wave of one–ninth its frequency as its
duty cycle is varied.
1.0
0.75
0.5
0.25
0
–0.25
–0.5
–0.75
–1.0
Input
Output
Switching Frequency =
9X Modulation Frequency
+1
0
–1
0°
90°
180°
270°
360° 420°
Figure 1. Switching Amplifier Basic Waveforms
The concept of switching amplifiers has been around for
about 50 years but they were impractical before the advent
of complementary TMOS power MOSFETs. Vacuum tubes
were fast enough but they were rather poor switches. A
totem pole circuit with supply voltages of ±250 volts would
drop about 50 volts when switching a current of 200
milliamps. The efficiency of a tube switching amp could
therefore not exceed 80%. The transformer needed to
match the high plate impedance to the low impedance
speaker filter was impractical as well.
This document may contain references to devices which are no
longer offered. Please contact your ON Semiconductor represen-
tative for information on possible replacement devices.
With the introduction of complementary bipolar power
transistors in the late 1960s, switching amplifiers became
theoretically practical. At low frequencies, bipolar transistors
have switching efficiencies of 99% and will directly drive
a low impedance speaker filter. The requirement for
switching frequencies above 100 kHz resulted in excessive
losses however. Bipolar drive circuitry was also complex
because of its large base current requirement.
With the advent of complementary (voltage/current
ratings) TMOS power MOSFETs, gate drive circuitry has
been simplified. These MOS devices are very efficient as
switches and they can operate at higher frequencies.
A block diagram of the amplifier is shown in Figure 2.
An output switch connects either +44 or –44 volts to the
input of the low pass filter. This switch operates at a carrier
frequency of 120 kHz. Its duty cycle can vary from 5% to
95% which allows the speaker voltage to reach 90% of
either the positive or negative supplies. The filter has a
response in the audio frequency range that is as flat as
possible, with high attenuation of the carrier frequency and
its harmonics. A 0.05 ohm current sense resistor (R27) is
used in the ground return of the filter and speaker to provide
short circuit protection.
The negative feedback loop is closed before the filter to
prevent instabilities. Feedback cannot be taken from the
speaker because of the phase shift of the output filter, which
varies from 0° at dc to nearly 360° at 120 kHz. Since the
filter is linear, feedback may be taken from the filter input,
which has no phase shift. Unfortunately, this point is a high
frequency square wave which must be integrated to
determine its average voltage. The input is mixed with the
square wave output by resistors R4 and R5 shown in Figure
2. The resultant signal is integrated, which accurately
simulates the effect of the output filter. The output of the
integrator will be zero only if the filter input is an accurate
inverted reproduction of the amplifier input. If the output
is higher or lower than desired, the integrator will generate
a negative or positive error voltage. This error voltage is
applied to the input of the switch controller, which makes
the required correction. The integrator introduces a 90°
phase shift at high frequencies which leaves a phase margin
of nearly 90°.
© Semiconductor Components Industries, LLC, 2002
August, 2002 – Rev. 3
1
Publication Order Number:
AN1042/D
Free Datasheet http://www.datasheet4u.com/
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AN1042/D
wave clock is fed to a pair of CMOS monostables each of
which produces a 250 nanosecond pulse. Only timing
resistors are used and internal parasitics serve as the timing
capacitance. One monostable produces a pulse on the
positive transition of the square wave, and the other
produces a pulse on the negative transition. These short
pulses are connected to the control inputs of two CMOS
analog switches. When the 120 kHz square wave goes
positive, the upper CMOS switch turns on and the common
terminal is switches to + 5 volts. When the 120 kHz square
wave goes negative, the bottom CMOS switch turns on and
the common terminal is switched to –5 volts.
Since the drive signal from U1D is fed through R9, it will
be overridden if either of the CMOS switches is on. If the
error voltage to U1D is out of limits, its output will be
locked up at either +5 or –5 volts. The CMOS switches will
then act to insure either short negative or positive pulses to
the input of U1C. U1C is a comparator used as an inverting
buffer between the CMOS switches and the small signal
TMOS drivers. These devices have low input capacitance
and low output impedance.
The drive signal is fed through R12 to the gates of Q1 and
Q2. They function as a low impedance inverting buffer to
drive the output stage. Decoupling networks isolate the
sources of Q1 and Q2 from the ±5 volt supplies. This
prevents the disruption of other circuitry by the large
current spikes needed to drive the output stages. Note that
the feedback path from R5 to the output experiences 5
polarity inversions. They are U2C, U1D, U1C, Q1–Q2 and
Q3–Q4. An odd number of inversions is required to make
the overall feedback negative.
The current limiting circuitry is shown in Figure 8. R27,
a 0.05 ohm noninductive resistor, senses the ground current
in the output filter and speaker. The voltage across this
resistor is amplifier by op amp U2D. R28 and R29 set the
gain of U2D at 10. C11 rolls off the response above 300 kHz.
The level at the output of U2D is –0.5 volt per amp of output
current. The output of U2D is applied through R8 to the
error amp for filter resistance compensation as shown in
Figure 6. For every amp drawn by the speaker, the output
voltage is increased by about 0.1 volt. This compensates for
the loss in the filter and current sensing resistor. The
lowered output impedance at low frequencies improves
speaker damping.
The amplified current signal at the output of U2D is also
routed to the noninverting inputs of U2A and U2B. These
op amps are the current limiters. U2A limits negative
current and U2B limits positive current. Only U2A will be
described since U2B operates in an identical manner. R19
and R21 form a voltage divider with an output of 2.5 volts.
This voltage is applied to the inverting input of U2A. When
the non–inverting input of U2A is more positive than 2.5
volts, the speaker current is greater than –5 amps. In that
case, the output of U2A will rise towards +5 volts. This
output coupled through CR1 takes over control of the error
voltage buss. A voltage between ±2 volts is rapidly reached
R25
C12
R23
–
U2A
+
(Op Amp)
CR1
+5
R21
R19
R20
–5
R22
(Op Amp)
–
U2B
+
CR2
R24
R26 C13
–5 V
R29
C11
U2D
(Op Amp)
R28
Current
Limit
R30
U1 Pin 8
R31
U1 Pin 9
R32
U2 Pin 8
R33
U2 Pin 9
Current
Compensation
From Current
Sense Resistor
Figure 8. Schematic of Current Limiting and
Current Sense Amplifier
at the output of CR1 to limit the current at –5 amps. Note
that U2B has +5 volts for its output at this time and CR2 is
reverse biased. R23 limits the low frequency gain of U2A
to 45. R25 in conjunction with C12 limits the high
frequency gain. If the output current exceeds –5 amps by
as little as 0.1 amp, the output voltage can be reduced to
zero from full voltage.
The resistor–capacitor combination of R25 and C12
form a lag compensation filter. They are necessary because
the output inductors introduce a 90° lag in output current
near 1 kHz when the output is shorted. The values chosen
for the lag filter are a compromise between speed of
response and stability under short circuit conditions. An
overcurrent of 0.1 amp requires about 50 microseconds to
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